Electric vehicle traction motor control

ABSTRACT

An electric vehicle traction motor control circuit includes a main thyristor in series with the motor armature. A second thyristor is connected in series with a commutating capacitor across the main thyristor which can be switched off by firing the second thyristor so that the motor current is diverted into the commutating capacitor. A third thyristor is connected in series with an inductor across the commutating capacitor and serves when fired to reverse the voltage residing on the commutating capacitor following commutating. The reversed voltage on the capacitor is detected and is used to control upper and lower current limit signals set by a driver operable control so that the capacitor is not called upon to receive a large current when it is in a low voltage condition.

This invention relates to an electric vehicle motor control of the general kind in which motor current is controlled by periodically making and breaking a main current path to the motor when the motor current falls below a lower limit and rises above an upper limit respectively, the limits being variable under the control of the vehicle driver.

In one known arrangement the main current path is constituted by a first thyristor connected in series with the motor between a pair of supply rails. To turn off the main thyristor it is required to divert all the motor current through a communicating path which consists of a second thyristor, first inductor and a capacitor in series across the main thyristor. A third thyristor is connected in series with a second inductor across the capacitor and is arranged when fired to reverse the voltage across the capacitor.

With such arrangements various firing orders have been proposed for the thyristors. One possibility is to fire the main thyristor and the third thyristor simultaneously when the lower limit is reached and to fire the second thyristor when the upper limit is reached. With this arrangement problems arise if the upper limit is not reached, because the reversed charge on the capacitor may leak away so that when the second thyristor is eventually fired, proper commutation does not occur. To overcome this, and other linked problems, the control circuits associated with the thyristor, have in the past been relatively complex, to provide minimum and maximum on and off times for the main thyristor.

Other approaches to this problem have also been proposed but all put unwanted constraints on the control system.

It is an object of the present invention to provide an improved control circuit in which the problems referred to are substantially avoided.

An electric vehicle traction motor control circuit in accordance with the invention comprises a main thyristor in series with the motor armature between a pair of supply rails, a second thyristor arranged on firing to divert current from the main thyristor into a commutating capacitor, a third thyristor connected in series with an inductor across the commutating capacitor and arranged on firing to reverse the voltage on the commutating capacitor, driver operable control means for determining upper and lower motor current limits, means sensitive to the magnitude of a voltage stored on the commutating capacitor for varying said motor current limits independently of said driver operable control means first firing control means for firing the main thyristor when the actual motor current is less than said lower limit, second firing control means for firing the second thyristor when the actual motor current exceeds the upper limit and third firing control means for the third thyristor.

An example of the invention is shown in the accompanying drawings in which:

FIG. 1 is a circuit diagram of a thyristor chopper and motor circuit;

FIG. 2 is a circuit diagram of a commutating capacitor voltage sensing circuit;

FIG. 3 is a circuit diagram of a current comparator circuit and the firing circuit for one of the thyristors of the chopper circuit of FIG. 1;

FIGS. 4 and 5 are circuit diagrams of firing circuits for the two other thyristors of FIG. 1 respectively; and

FIGS. 6 and 7 are circuit diagrams of other logic circuit elements included in the control circuit.

Referring firstly to FIG. 1 the motor in this case has an armature winding 10 and a field winding 11. A contact F connects one side of the armature winding 10 to a positive supply rail 12 (at about 200 V relative to a rail 13) and a contact R connects the other side of the armature 10 to the rail 12. The two sides of the armature 10 are also connected to two fixed contacts of a change-over switch F/RB, the common contact of which is connected via an inductor 14 the field winding 11, a main thyristor SCR1 and a main fuse 15 in series to the rail 13. Four diodes D₁ to D₄ connect the two sides of the armature 10 to the rails 12, 13, with the diode D₁ having its anode connected to the same side of the armature as the contact F and its cathode connected to the rail 12, the diode D₂ having its cathode connected to the anode of the diode D₁ and its anode connected to the rail 13, the diode D₃ having its anode connected to the other side of the armature 10 and its cathode connected to the rail 12 and the diode D₄ having its cathode connected to the anode of the diode D₃ and its anode connected by a brake current fuse 16 to the rail 13. A further diode D₅ has its anode connected to the anode of the thyristor SCR1 and its cathode connected to the rail 12. A sixth diode D₆ has its cathode connected to the common contact of the switch F/RB and its anode connected to the rail 13.

For commutating the current through the thyristor SCR1 there is a second thyristor SCR2 connected in series with a saturable core inductor 17, a fuse 18 and a commutating capacitor 19 between the anode of the thyristor SCR1 and the rail 13. A third thyristor SCR3 is connected in series with an inductor 20 across the capacitor 19.

For normal forward motoring the contact F is closed and the switch F/RB is closed to the right by contactors driven by circuits which do not form part of the present invention and an understanding of which is not necessary for an understanding of the present invention. When the main thyristor is conductive current flows through contact F, "forwardly" through the armature through the switch F/RB, the inductor 14, the field winding 11, the thyristor SCR1 and the main fuse 15. When the thyristor SCR2 is first (assuming that there is a negative voltage on the "upper" plate of the capacitor 19 at this stage), the current referred to is diverted from the main thyristor SCR1 into the capacitor 19 via the inductor 17. This causes the thyristor SCR1 to turn off. The diverted current causes the capacitor 19 to charge up until the voltage on the upper plate of the capacitor 19 is the same as that at the anode of the thyristor SCR1. At this stage the inductor 17 continues to cause current to flow into the capacitor 19 and the inductor 14 acts to maintain current in the armature and field windings via the "freewheel" diode D₅. Inductor 17 and capacitor 19 act as a resonant circuit but when the voltage on capacitor 19 reaches its peak, (i.e. when the current in the inductor 17 falls to zero) the thyristor SCR2 turns off and the charge is then held on the capacitor 19. The armature and field winding current decays away.

Firing of the thyristor SCR3 causes the inductor 20 to be connected across the charged capacitor 19. The capacitor 19 thus discharges into the inductor 20, current continuing to flow until a peak negative voltage is attained, which peak is held until the next commutation is required.

For reverse running contact F is opened, contact R is closed and switch F/RB is closed to the left. For regenerative braking when the motor is running forwardly both contacts F and R are opened and switch F/RB is closed to the left. Current induced in the armature, then flows through the switch F/RB, the inductor 14, the field winding 11, the thyristor SCR1, the fuses 15 and 16 and the diode D₄.

FIG. 1 also shows an armature current sensing arrangement comprising a ferromagnetic loop 21 surrounding one of the conductors leading to the armature winding 10, a Hall effect device 22 in a gap in this loop and a differential amplifier A₁ with its inputs connected by resistor R₁ and R₂ to the terminals of the device 22. A feedback resistor R₃ connects the inverting input terminal of the amplifier A₁ to its output terminal and a bias resistor R₄ connects the non-inverting terminal to earth. A resistor and thermistor network 23 is provided at the output of the amplifier A₁ to provide temperature compensation.

A circuit is also provided to detect the instant at which the voltage on the inductor 17 reverses and the diode D₅ starts to conduct during commutation. In the example shown this circuit includes a diode D₇, a resistor R₅ and the light emitting diode of an opto-coupler O₁ connected in series across the inductor 17. The transistor of the opto-coupler is included in a "ready" circuit (not shown in detail) which produces an output pulse at a terminal 24 for as long as there is a reversed voltage on the inductor 17. The "ready" pulse may alternatively be generated by means of a current detector on the conductor associated with the freewheel diode D₅. Such detector may be a current transformer or another Hall effect type detector, the latter being preferred.

Turning now to FIG. 2 there is shown a circuit which is used to monitor the voltage on the capacitor 19 after firing of the thyristor SCR3. This circuit includes a pnp transistor Q₁ which has its collector connected to the rail 13 and its base connected by a high ohmic value resistor R₅ to the capacitor 19. The emitter of the transistor Q₁, is connected via a current limiting resistor R₆ and the light emitting diode of an opto-coupler O₂ to a rail 25. A resistor R₇ is connected between the base of the transistor Q₁ and the rail 25 and a diode D₈ has its cathode connected to the rail 25 and its anode connected to the base of transistor Q₁ to provide protection of the transistor Q₁ in the period whilst the capacitor voltage is positive following commutation. The rail 25 has a zener diode regulated supply from the rail 12. To this end a resistor R₉ connects the rail 12 to the rail 25 and a zener diode ZD₁ has its cathode connected to the rail 25 and its anode connected to the rail 13, a smoothing capacitor C₁.

The photo-transistor of the opto-coupler O₂ has its collector connected to a +15 V supply rail 26 associated with a ground rail 27 and a -15 V supply rail of a power supply which is isolated from the main traction power supply rails 12, 13. The emitter of this photo-transistor is connected by a resistor R₁₀ to the rail 27 and a resistor R₁₁ connects its base to its emitter. The photo-transistor of another opto-coupler O₃ is likewise connected with resistors R₁₂ and and R₁₃. The emitters of these two transistors are also connected by resistors R₁₄ and R₁₅ respectively to the inverting and non-inverting input terminals of an operational amplifier A₁. A resistor R₁₆ connects the non-inverting input terminal of the amplifier A₁ to the rail 27 and the output terminal of the amplifier A₁ is connected to the cathode of a diode D₉ with its anode connected to the base of an pnp transistor Q₂ with a potentiometer R₁₇ connecting the emitter of the transistor Q₂ to the rail 27. The collector of the transistor Q₂ is connected via light emitting diode of the opto-coupler O₃ to the rail 28. A resistor R₁₈ connects the base of the transistor Q₂ to the rail 27. The emitter of the transistor Q₂ is connected by a resistor R₁₉ and a capacitor C₂ in parallel to the inverting input terminal of the amplifier A₁.

The circuit described provides a voltage across the potentiometer R₁₇ which is substantially linearly related to the voltage across the capacitor 19 (except when this latter voltage is positive following commutation). The use of the opto-isolator O₃ in the feedback loop of the amplifier A₁ provides compensation for the non-linearity and variations of gain with temperature of the opto-isolator O₂, assuming the two opto-isolators O₂ and O₃ to be reasonably matched.

The slider of the potentiometer R₁₇ is connected both to a sample and hold circuit based on operational amplifiers A₂ and A₃ and to a threshold voltage detector based on an operational amplifier A₄. The sample and hold circuit includes an n-channel field effect transistor Q₃ with its drain connected to the output terminal of the amplifier A₂ and its source connected to one side of a capacitor C₃ which has its other side connected to the rail 27. The inverting input terminal of the amplifier A₂ is connected by a resistor R₁₉ to the slider of the potentiometer R₁₇ and its non-inverting input terminal is connected by a resistor R₂₀ to the rail 27 and by a resistor R₂₁ to the rail 28. A feedback resistor R₂₂ connects the output terminal of the amplifier A₂ to the inverting input terminal thereof so that the amplifier A₂ acts as a linear inverting amplifier.

A bias resistor R₂₃ connects the output terminal of the amplifier A₂ to the gate of the field effect transistor Q₃ which is also connected to the anode of a diode D₁₀. The cathode of the diode D₁₀ is connected by a resistor R₂₄ to the collector of an npn transistor Q₄ which has its emitter connected to the rail 28 and its collector connected by a resistor R₂₅ to the rail 26. The base of the transistor Q₄ is connected to the common point of two resistors R₂₆, R₂₇ in series between the collector of a pnp transistor Q₅ and the rail 28. The emitter of the transistor is connected to the rail 26 and it is biased off by a resistor R₂₈ connected between the rail 26 and the base of the transistor Q₄ which is also connected by a resistor R₂₉ to a terminal 29 (see FIG. 7).

Whilst the terminal 29 is at a voltage close to that on the rail 26, the transistors Q₄ and Q₅ are off and the field effect transistor Q₃ is non-conductive. When the voltage on the terminal 29 falls as will be explained hereinafter, the transistor Q₄ and Q₅ turn on and the field effect transistor Q₃ assumes a low resistance state, allowing the capacitor C₃ to charge or discharge rapidly to the voltage then existing at the output terminal of the amplifier A₂.

The amplifier A₃ is connected as a voltage follower to provide a very high input impedance so as not to discharge the capacitor C₃. A resistor R₃₀ connects the capacitor C₃ to the non-inverting input terminal of the amplifier A₃, a resistor R₃₁ connecting the output terminal of the amplifier A₃ to its inverting input terminal. The output terminal of the amplifier A₃ is also connected to the cathode of a diode D₁₁, of which is connected by a normally closed relay contact to a terminal 30 (FIG. 3).

The amplifier A₄ is connected as a comparator with hysteresis. Its inverting input terminal is connected by a resistor R₃₂ to the slider of the potentiometer R₁₇, and its non-inverting input terminal is connected to the common point of a pair of resistors R₃₃, R₃₄ in series between the rails 27, 28 and also, by a resistor R₃₅, to its output terminal which is connected to a terminal 31.

Turning now to FIG. 3, there is shown schematically the arrangement in which the signals from the current transducer circuit of FIG. 1 and the capacitor voltage circuit of FIG. 2 are used. The current signal is applied via a resistor R₄₁ to the inverting input terminal of an operational amplifier A₅ connected as a comparator with hysteresis. The required hysteresis is obtained by means of resistors R₄₂, R₄₃ connected in series between the output terminal of the amplifier A₅ and the rail 27 with their common point connected to the non-inverting input terminal of the amplifier A₅. The inverting input terminal of the amplifier A₅ is connected by a resistor R₄₄ to a motor current demand signal generating circuit 32, of which the output stage is shown. It receives input signals from a speed transducer circuit 33 and from accelerator and brake pedal operated potentiometers 34 and 35 and produces a d.c. output signal representative of the desired average motor current varying in accordance with a complex function of speed and pedal depression. Details of a similar circuit arrangement may be found in prior U.K. Patent Application No. 8364/75.

The output stage is constituted by an operational amplifier A₁₀₀ which operates either in inverting or non-inverting mode according to whether a transistor Q₁₀₀ is on or not. The inverting input terminal of the amplifier A₁₀₀ is connected by a resistor R₁₅₀ to a point in the circuit 32 and two resistors R₁₅₁, R₁₅₂ of the same value connect the same point to the non-inverting input terminal. A resistor R₁₅₃ combined with a capacitor C₁₀₀ in series connect the output terminal of the amplifier A₁₀₀ to the earth rail 27. A resistor R₁₅₄ connects the junction of the resistor R₁₅₃ with the capacitor C₁₀₀ to the anode of a diode D₁₀₀, the cathode of which is connected to the base of a transistor Q₁₀₁ connected as an emitter follower. Two resistors R₁₅₅, R₁₅₆ in series connect the emitter of the transistor Q₁₀₁ to the rail 27 with the junction of these resistors connected to the inverting input terminal of the amplifier A₁₀₀ by a feedback resistor R₁₅₇. The emitter of the transistor Q₁₀₁ is connected by the resistor R₄₄ to the inverting input terminal of the amplifier A₅.

The terminal 30 is connected to the base of the emitter follower transistor Q₁₀₁. If the output of the amplifier A₅ is high the voltage at its non-inverting input terminal is at a fixed positive level which must be exceeded by the voltage at its inverting input terminal before the amplifier output can go negative. Similarly when the output of the amplifier A₅ is low, there is a fixed negative voltage applied to the non-inverting input terminal of the amplifier A₅. The signals applied via resistors R₄₁ and R₄₄ are of opposite polarity and these resistors may be regarded as forming a potential divider so that for a given demand signal the output of the amplifier A₅ will go low when an upper current limit is exceeded and will go high when the motor current is less than a low current limit, both limits being variable by the driver using the pedals 34, 35.

The effect of the capacitor voltage signal applied via terminal 30 is to decrease the level to which the current demand signal may rise if capacitor voltage is too low.

The output terminal of the amplifier A₅ is connected by a diode D₁₂ and a resistive potential divider R₄₆, R₄₇ to the base of an npn transistor Q₆ with its emitter grounded to the rail 27 and its collector connected by a resistor R₄₈ to the rail 26. The collector of the transistor Q₆ is connected to a terminal 36 (FIG. 4) and also to the cathode of a diode D₁₃ with its anode connected by a resistor R₄₉ to the rail 26. A diode D₁₄ connects the anode of the diode D₁₃ to a terminal 37 (FIG. 7). A zener diode ZD₂ has its cathode connected to the anode of the diodes D₁₃ and D₁₄ and its anode connected by a resistor R₅₀ to the rail 27. The resistor R₄₉ and the zener diode ZD₂ are chosen so that the zener diode ZD₂ does not conduct if either of the diodes D₁₃, D₁₄ is conducted, i.e. if the transistor Q₆ is on or the terminal 39 voltage is low.

An npn transistor Q₇ has its emitter connected to the rail 27 and its base connected to the anode of the zener diode ZD₂, its collector being connected by a resistor R₅₁ to the rail 26. The transistor Q₇ turns on whenever the zener diode ZD₂ is conducting.

The emitter of the transistor Q₇ is connected by a resistive potential divider R₅₂, R₅₃ to the base of an npn transistor Q₈ which controls an oscillator based on a unijunction transistor Q₉ and a capacitor C₄. The secondary base of the unijunction transistor Q₉ is connected by two resistors R₅₄, R₅₅ in series to the rail 27 and its primary base is connected by a resistor R₅₆ to the rail 26. There are two separate charging paths for the capacitor C₄ which determine the frequency of the oscillator. One such path is constituted by a resistor R₅₇ connected in series with the capacitor C₄ between the rails 26, 27. The other path is constituted by a resistor R₅₈, a diode D₁₅, a diode D₁₆ and a resistor R₅₉ in series across the resistor R₅₇, the total resistance of the resistors R₅₈, R₅₉ being significantly less than the resistance of the resistor R₅₇. A diode D₁₇ is connected across the series combination of the resistor R₅₉ and the diode D₁₆, with its polarity reversed relative to that of the diode D₁₆. The anode of the diode D₁₆ and the cathode of the diode D₁₇ are connected to the collector of the transistor Q₈. The anode of the diode D₁₅ is connected to the anode of a diode D₁₈, the cathode of which is connected to the anode of a thyristor SCR4 which has its cathode connected to the rail 27.

When the transistor Q₈ is conducting, the oscillator does not operate since the capacitor C₄ is held discharged. When the transistor Q₈ is turned off, i.e. when the transistor Q₇ turns on, the oscillator starts to run at a relatively high frequency until the thyristor SCR4 is fired as will be explained hereinafter. Thereafter, for as long as the transistor Q₈ is off, the oscillator will operate at a low frequency determined by the resistor R₅₇.

The common point of the resistors R₅₄, R₅₅ is connected to the base of a transistor Q₁₀ which has its emitter connected to the rail 27 and its collector connected by a resistor R₆₀ to the rail 26. The collector of the transistor Q₁₀ is connected to the TRIGGER terminal of an integrated circuit timer T₁ (shown as one half of a dual timer circuit type NE556 manufactured by signetics). The DISCHARGE and THRESHOLD terminals of this timer circuit are connected to the junction of a resistor R₆₁ and a capacitor C₅ in series between the rails 26 and 27. The CONTROL VOLTAGE terminal of the circuit is connected to the rail 27 by a capacitor C₆ and the RESET terminal is connected to the collector of an npn transistor Q₁₁ which has its emitter connected to the rail 27 and its collector connected by a resistor R₆₂ to the rail 26. The base of the transistor Q₁₁ is connected by a resistor R₆₃ to the rail 27 and by a resistor R₆₄ to the cathode of a diode D₁₉ having its anode connected to the collector of the transistor Q₇. The base of the transistor Q₁₁ is also connected by a resistor R₆₅ to the cathode of a diode D₂₀, the anode of which is connected to a terminal 40.

The OUTPUT terminal of the timer circuit T₁ is connected to a terminal 41 and thence to the gate of the main thyristor SCR1. This OUTPUT terminal is also connected by a normally closed override contact 42 and a resistor R₆₆ to gate of the thyristor SCR4, this gate being also connected by a resistor R₆₇ to the rail.

Assuming the voltage at terminal 40 is low and that at terminal 89 is high turning off of the transistor Q₆ by a negative transition in the output of the output of the current comparator operational amplifier A₅ causes transistor Q₇ to turn on which in turn causes the oscillator to commence oscillating at relatively high frequency, and, because, in these circumstances, the transistor Q₁₁ is turned off, the timer T₁ will be triggered the resistors R₅₈ and R₅₉ and the capacitor C₄ fixing the delay before such triggering at approximately 70 μS. The timer OUTPUT terminal now goes high for 20 μS, set by the resistor R₆₁ and the capacitor C₅. This output pulse fires the thyristors SCR1 and SCR4, the latter interrupting the previous charging path for the capacitor C₄. The resistor R₅₇ and the capacitor C₄ set the repetition frequency of the oscillator at approximately 200 Hz so that additional firing pulses are produced by the timer T₁ at this frequency, in case the current level in the thyristor SCR1 is insufficient for it to hold the thyristor SCR1 conductive when the output of the current comparator operational amplifier A₅ goes high (indicating that the actual current has reached the upper limit), transistor Q₆ turns on, turning off transistor Q₇ which turns on transistors Q₈ and Q₁₁. Transistor Q₁₁ holds the RESET terminal of the timer T₁ low so that it cannot be triggered and transistor Q₈ maintains the capacitor C₄ discharged so that the oscillator ceases to run.

The terminals 36, 39 and 40 provide input signals to the circuit of FIG. 4 which controls firing of the thyristor SCR2. A resistor R₇₀ connects the terminal 36 to the anode of a zener diode ZD3, the cathode of which is connected by a resistor R₇₁ to the rails 26 so that the zener diode ZD3 only conducts when the transistor Q₆ of FIG. 3 is on (i.e. when the current comparator operation amplifier A₅ output is high). The cathode of the zener diode ZD3 is connected to the base of a pnp transistor Q₁₂, the emitter of which is connected to the rail 26 and the collector of which is connected to the rail 27 by a resistor R₇₂. The collector of the transistor Q₁₂ is connected by a resistor R₇₃ and a capacitor C₆ in series to the base of a npn transistor Q₁₃ which has its emitter connected to the rail 27 and its collector connected by a resistor R₇₄ to the rail 26. The base of the transistor Q₁₃ is connected to the cathode of a protective diode D₂₂ which has its anode grounded to rail 27 and is also connected by a resistor R₇₅ to the rail 27 to bias the transistor Q₁₃ off.

It will be appreciated that when the transistor Q₁₂ turns on the transistor Q₁₃ will be turned on for as long as it takes the capacitor to charge up (the time constant R₇₃ C₆ being approximately 10 μS).

The collector of the transistor Q₁₂ is also connected by a resistor R₇₆ and a diode D₂₃ in series to one side of a capacitor C₇, the other side of which is connected to the rail 27. Said one side of said capacitor C₇ is connected to the emitter of a unijunction transistor Q₁₄, which has its secondary base connected by a resistor R₇₇ to the rail 27 and its primary base connected to the rail 26 by a resistor R₇₈. The secondary base of the unijunction transistor Q₁₄ is also connected by a resistor R₇₉ and a diode D₂₄ in series to the base of the transistor Q₁₃.

The resistor R₇₆, capacitor C₇ and unijunction transistor Q₁₄ form an oscillator, operating under the control of the transistor Q₁₂ at a frequency of about 3 KHz. This oscillator is arranged to be inhibited by an npn transistor Q₁₅ with its collector emitter connected across the capacitor C₇ and biased off by a resistor R₈₀ connected between its base and the rail 27. The base of the transistor Q₁₅ is also connected to the cathode of a thyristor SCR5 the anode of which is connected by a resistor R₈₁ to the collector of the transistor Q₁₂. The base of the transistor Q₁₅ is further connected to the collector of an npn transistor Q₁₆ having its emitter connected to the rail 27 and its base connected by a resistor R₈₂ to the rail 27. A diode D₂₄ has its cathode connected to the base of the transistor Q₁₆ and its anode connected by a resistor R₈₃ to the terminal 40. A diode D₂₅ connects the anode of the diode D₂₄ to the terminal 39 so that the transistor Q₁₆ can only turn on when the signals at both terminals 39 and 40 are high.

The thyristor SCR5 has its gate connected by a resistor R₈₄, a diode D₂₆ and a normal closed override contact 43 to the OUTPUT terminal to another integrated circuit timer T₂ (constituted by the other half of the NE556 circuit of time T₁). The INPUT terminal of the timer T₂ is connected to the emitter of the transistor Q₁₃ and its CONTROL VOLTAGE terminal is connected by a capacitor C₈ to the rail 27. The THRESHOLD and DISCHARGE terminals of the timer T₂ are both connected by a resistor R₈₅ to the rail 26 and by a capacitor C₉ to the rail 27, the resistor R₈₅ and the capacitor C₉ setting the on-time of the monostable multivibrator constituted by the timer T₂ and its associated components at about 20 μS. The OUTPUT terminal of the timer T₂ is also connected to the gate of the second thyristor SCR2 (FIG. 1) and to a terminal 44.

The RESET terminal of the timer T₂ is connected by a resistor R₈₆ to the rail 26 and by the collector-emitter path of an npn transistor Q₁₇ to the rail 27. The base of the transistor Q₁₇ is connected by a resistor R₈₇ to the rail 27 and by a resistor R₈₈ to the anode of a zener diode ZD4 having its cathode connected to the cathode of three diodes D₂₆, D₂₇, and D₂₈ respectively. The anodes of the diodes D₂₆, D₂₇, and D₂₈ are connected to three terminals 45, 46 and 46 see FIG. 5 respectively, so that a high voltage signal at any of these terminals will turn on the transistor Q₁₇ and reset the timer T₂. A capacitor C₁₀ is connected between the cathode of the zener diode ZD4 and the rail 27.

Provided that the signal voltages at the terminals 45, 46 and 47 are all low when the transistor Q₁₂ is turned on by output of the current comparator operational amplifier A₅ going low, the timer T₂ will be triggered immediately and the oscillator around the unijunction transistor Q₁₄ starts to run providing further triggering pulses to the timer T₂. The first output pulse from the timer T₂ turns on the thyristor SCR5 which, unless the transistor Q₁₆ is on, turns on the transistor Q₁₅ on, thereby holding the capacitor C₇ discharged and stopping the oscillator from running.

A normally open contact 48 connects the terminal 36 via a resistor R₈₉ and a diode D₂₉ to the emitter of the unijunction transistor Q₁₄, so that when the contact 48 is closed (when a "creep" condition has been selected) the oscillator runs at a significantly lower frequency of say 400 Hz when the output of the current comparator operational amplifier A₅ is low. In this condition the transistor Q₁₂ is off so that firing of the thyristor SCR5 does not provide current to turn on the transistor Q₁₅ and stop the oscillator.

Turning now to FIG. 5 the OUTPUT terminal of the timer T₂ (FIG. 4) is connected via the terminal 44 to through two resistors R₉₀, R₉₁ in series to the rail 27. The common point of these resistors is connected to the base of an npn transistor Q₁₈, the emitter of which is grounded to the rail 27. The collector of the transistor Q₁₈ is connected by a resistor R₉₂ to the rail 26 and is also connected to the TRIGGER terminal of another timer integrated circuit T₃ which is connected as a monostable multivibrator with a pulse duration of about 2.5 mS. The THRESHOLD and DISCHARGE terminals of the timer T₃ are the rail 27 by a capacitor C₁₁ and to the rail 26 by a resistor R₉₃ and variable reset R₉₄ in series. A capacitor C₁₂ connects the CONTROL VOLTAGE terminal of the timer T₃ to the rail 26 and the RESET terminal is connected to a terminal R₁. The OUTPUT terminal is connected to a terminal 49 (FIGS. 6 and 7).

The output terminal of the timer T₃ is also connected via a capacitor C₁₃ to one end of a resistor R₉₅ the other end of which is connected to the rail 26. A diode D₂₉ has its cathode connected to the rail 26 and its anode connected to the junction between the capacitor C₁₃ and the resistor R₉₅ so that a negative going pulse is produced at this junction when the timer T₃ resets. This junction is also connected to the TRIGGER terminal of another timer T₄, again connected as a monostable multivibrator with a pulse duration of about 20 μS. The THRESHOLD and DISCHARGE terminals of the timer T₄ are connected to the rail 26 by a resistor R₉₆ and to the rail 26 by a capacitor C₁₄. A capacitor C₁₅ connects the CONTROL VOLTAGE terminal of the timer T₄ to the rail 27 and its RESET terminal is connected to a terminal R₂. The output terminal of the timer T₄ is connected via the terminal 45 to the gate of the thyristor SCR₃ and also to the reset circuit of the timer T₂ (FIG. 4).

The OUTPUT terminal of the timer T₄ is also connected by two resistors R₉₈, R₉₉ in series to the rail 27, the junction of these resistors being connected to the base of an npn transistor Q₁₉ which has its emitter connected to the rail 27. The collector of the transistor Q₁₉ is connected by a load resistor R₁₀₀ to the rail 26 and is also connected to the TRIGGER terminal of yet another timer T₅ connected as a monostable multivibrator with a pulse duration of 300 μS. The THRESHOLD and DISCHARGE terminals of the timer T₅ are connected to the rail 27 by a capacitor C₁₆ and to the rail 26 by a resistor R₁₀₁ and a variable resistor R₁₀₂ in series. The CONTROL VOLTAGE terminal of the timer T₅ is connected by a capacitor C₁₇ to the rail 27 and its RESET terminal is connected directly to the rail 26.

A capacitor C₁₈ connects the OUTPUT terminal of the timer T₅, which terminal is also connected to the terminal 46, to the TRIGGER terminal of a further timer circuit T₆ connected as a monostable multivibrator with a pulse length of 100 μS. The THRESHOLD and DISCHARGE terminals of the timer T₆ are connected to the rail 27 by a capacitor C₁₉ and to the rail 26 by a resistor R₁₀₃. The CONTROL VOLTAGE terminal of the timer T₆ is connected by a capacitor C₂₀ to the rail 26 and its RESET terminal is connected directly to the rail 26. The TRIGGER terminal is connected by a resistor R₁₀₄ and a diode D₃₀ to the rail 27 so as to convert the trailing edge of the output pulse of the timer T₅ into a negative-going triggering inpulse for the timer T₆. The output terminal of the timer T₆ is connected to the terminal 47.

Turning now to FIG. 6, there is shown a simple circuit which converts the low signal which exists at terminal 31 when the capacitor 19 voltage is too low into a high level signal for use in the circuits of FIGS. 3 and 4. This circuit includes a resistor R₁₁₀ connecting the anode of a diode D₃₁ to the rail 26. The terminal 31 is connected to the anode of this diode by a normally closed contact which is energised if it is required to override the Vlow control signal. The cathode of the diode D₃₁ is connected by a resistor R₁₁₁ to the base of an npn transistor Q₂₀ which base is also connected by a resistor R₁₁₁ to the rail 27. The transistor Q₂₀ has its emitter connected to the rail 27 and its collector connected to the terminal 40 and by a resistor R₁₁₃ to the rail 26.

The R₂ reset signal for resetting the timer T₄ is generated by inverting a demand cutoff signal generated under certain circumstances during change overs between forward and reverse motoring and braking. This particular feature is not essential to the present invention and the derivation of the demand cutoff signal will not be described herein. The demand cutoff terminal 50 is connected by a resistor R₁₁₄ to the base of an npn transistor Q₂₁ which has its emitter connected to the rail 27. A resistor R₁₁₅ connects the base of the transistor Q₂₁ to the rail 27 and a resistor R₁₁₆ connects the collector of the transistor Q₂₁ to the rail 26, which collector is also connected to the terminal R₂ and to the cathode of a diode D₃₂. The anode of the diode D₃₂ is connected by a capacitor C to the rail 26 and also to the anodes of two diodes D₃₃, D₃₄ and a zener diode ZD5. The cathode of the diode D₃₃ is connected to terminal 49 and that of the zener diode ZD5 is connected by two resistors R₁₁₇ and R₁₁₈ in series to the rail 26. The junction of this resistor is connected to the base of a pnp transistor Q₂₂ with its emitter connected to the rail 26 and its collector connected by a resistor R₁₁₉ to the rail. The cathode of the diode D₃₄ is connected to the collector of an npn transistor Q23 with its emitter grounded to rail 27 and its collector connected by a resistor R₁₂₀ to the rail 27. The base of the transistor Q₂₃ is connected by resistor R₁₂₁ to the rail 27 and also to the anode of a zener diode ZD6 the cathode of which is connected to the terminal 24 and, by a resistor R₁₂₂ to the rail 26. This transistor Q₂₂ can be turned on by a positive going signal at the terminal 24 or at the terminal 50 or a negative going signal at the terminal 49.

The collector of the transistor Q₂₂ is connected to the TRIGGER terminal of another timer circuit T₇ connected as a monostable multivibrator with a pulse duration of 100 μS. The THRESHOLD and DISCHARGE terminals of the timer T₇ are connected by a capacitor C₂₂ to the rail 27 and by a resistor R₁₂₃ to the rail 26. The CONTROL VOLTAGE terminal is connected by a capacitor C₂₃ to the rail 27 and the RESET terminal is connected directly to the rail 26.

The OUTPUT terminal of the timer C₇ is coupled by a capacitor C₂₄ to the TRIGGER terminal of another timer T₈. This TRIGGER terminal is connected to the rail 26 by a resistor R₁₂₄ and a diode D₃₅ in parallel so that the timer T₈ is triggered by the falling edge of the output pulse of the timer T₇. The timer T₈ is connected as a monostable multivibrator with a pulse duration of 50 μS. Its THRESHOLD and DISCHARGE terminals are connected to the rail 27 by a capacitor C₂₅ and to the rail 26 by a resistor R₁₂₅. The CONTROL VOLTAGE terminal of the timer T₈ is connected to the rail 27 by a capacitor C₂₆. The RESET terminal of the timer T₈ is connected to the collector of an npn transistor Q₂₄ which has its emitter grounded to rail 27 and its collector connected to the rail 26 by a resistor R₁₂₆. The base of the transistor Q₂₄ is connected to the collector of an npn transistor Q₂₅, the emitter of which is connected to the rail 27 and the collector of which is connected by a resistor R₁₂₇ to the rail 26. The base of the transistor Q₂₅ is connected to the junction of two resistors R₁₂₈, R₁₂₉ connected in series between the OUTPUT terminal of the timer T₈ and the rail 27 and also connected by a resistor R₁₃₀ to the cathode of a diode D₃₆ with its anode connected to the collector of the transistor Q₂₃.

The OUTPUT terminal of the timer T₈ is also connected by two resistors Rhd 131 and R₁₃₂ in series to the rail 27, with the junction of these resistors connected to the base of an npn transistor Q₂₆. The emitter of the transistor Q₂₆ is connected to the rail 27 and its collector is connected via a normally closed override contact 51 to the R₁ terminal and also by a resistor R₁₃₃, to the rail 26.

The timer T₇ is triggered when the transistor Q₂₂ turns off either as a result of the signal at terminal 49 going high (caused by triggering of timer T₃) or as a result of the signal at terminal 24 going low. At the end of the output pulse from the timer T₇, the timer T₈ is triggered, provided to signal at terminal 24 remains low. Once triggered the timer T₈ is not reset if the signal at terminal 24 goes high.

Turning finally to FIG. 7, two simple logic circuits are shown. Firstly there is an npn transistor Q₂₇ controlling the signal at the terminal 29 in accordance with the signals at the terminals 46 and 49. The emitter of the transistor Q₂₇ is connected to the rail 27 and its base is connected to the same rail by a resistor R₁₃₅. This base is also connected to the anode of a zener diode ZD7, the cathode of which is connected by a resistor R₁₃₆ to the cathodes of two diodes D₃₇ and D₃₈, which have their anodes connected to the terminals 46 and 49 respectively. A capacitor C₂₇ is connected between the cathodes of these diodes and the rail 27. The collector of the transistor Q₂₇ is connected to the terminal 29 and by a resistor R₁₃₇ to the rail 26. It will be appreciated that the signal at the terminal 29 goes high to cause the field effect transistor Q₃ (FIG. 2) to become conductive, only when the signals at terminals 46 and 49 are both low.

The remainder of FIG. 7 controls the signal at terminal 37 in accordance with the signals at terminals 46, 47 and 49. An npn transistor Q₂₈ has its emitter connected to the rail 27 and its base connected to the same rail by a resistor R₁₄₀. This base is also connected to the anode of a zener diode ZD8, the cathode of which is connected by a resistor R₁₄₁ to the cathodes of three diodes D₃₉, D₄₀ and D₄₁ with their anodes connected to the terminals 46, 47 and 49 respectively. A capacitor C₂₈ is connected between the cathodes of these diodes and the rail 27. The collector of the transistor Q₂₈ is connected to the terminal 37 and also, by a resistor R₁₄₂ to the rail 26.

In normal running the sequence of operations is as follows:

The thyristor SCR1 is fired when the output of the current comparator operational amplifier A₅ goes low provided that the signal at terminal 39 is high (i.e. There ae currently no pulses being produced by timers T₃, T₅ and T₆) and the signal at terminal 40 is low (i.e. there is an adequate charge for commutation on the capacitor 19).

When the output of the amplifier A₅ goes high the timer T₁ is held reset (it will not still be producing its output pulse) via transistor Q₁₁ turning on when transistor Q₆ turns on. Also, transistor Q₁₂ turns on, causing transistor Q₁₃ to turn on momentarily and triggering the timer T₂ provided that the signals at terminals 45, 46 and 47 are all low (i.e. provided none of timers T₅, T₆ or T₇ is producing a pulse). This causes firing of the thyristor SCR2 to commutate the main thyristor SCR1 current.

When the thyristor SCR2 ceases to conduct (indicating completion of commutation) the signal at the terminal 24 goes low, so that, provided the signal at terminal 49 is high (which condition exists for 2.5 mS following firing of the thyristor SCR2 because of timer T₃), the timer T₇ is triggered. At the end of the timer T₇ output pulse timer T₈ is fired (assuming that the signal at the terminal 24 has not meanwhile gone high) and the output of the timer T₈ turns on the transistor Q₂₆ and thereby resets the timer T₃. Such resetting of the timer T₃ causes triggering of the timer T₄ which fires the thyristor SCR3. In the event of the reset pulse at R₁ being "missed" for any reason, the timer T₃ resets and triggers the timer T₄ after its full 2.5 mS delay.

Firing of the thyristor SCR3, causes triggering of the timer T₅ which triggers the timer T₆ after 300 μS, i.e. sufficient time for the voltage on the capacitor 19 to reverse and settle. During the 100 μS pulse duration of the timer, T₆, the f.e.t. Q₃ is conductive and the voltage on the capacitor 19 is sampled so that the signal at terminal 30 is adjusted to set the upper and lower current limits for the next cycle. Only when the signal at the terminal 47 goes low at the end of this 100 μS does the signal at the terminal 37 go high again to allow the next cycle to start.

Firing of thyristor SCR1 and SCR2 is prevented if the signal at terminal 31 goes low, but this can be overridden if required, for test purposes.

In the creep mode contact 48 is closed and the thyristors SCR2 and SCR3 are fired alternately whilst the signal at terminal 36 is high (i.e., after the motor current falls below its lower limit) so that some current can flow in the motor when the signal at terminal 36 goes low such alternate firing ceases until the motor current falls again below its lower limit. 

I claim:
 1. An electric vehicle traction motor control circuit comprising a main thyristor in series with the motor armature between a pair of supply rails, a second thyristor arranged on firing to divert current from the main thyristor into a commutating capacitor, a third thyristor connected in series with an inductor across the commutating capacitor and arranged on firing to reverse the voltage on the commutating capacitor, driver operable control means for determining upper and lower motor current limits, means sensitive to the magnitude of a voltage stored on the commutating capacitor for downwardly adjusting said motor current limits independently of said driver operable control means if said stored voltage is too low to ensure commutation, first firing control means for firing the main thyristor when the actual motor current is less than said lower limit, second firing control means for firing the second thyristor when the actual motor current exceeds the upper limit and third firing control means for the third thyristor.
 2. An electric vehicle traction motor control circuit as claimed in claim 1 in which said capacitor voltage sensitive means comprises means coupling said capacitor voltage to an amplifier, the output of which is related to the capacitor voltage, and a sample and hold circuit, whereby the output voltage of said amplifier is sampled periodically in synchronism with the operation of the motor control circuit.
 3. An electric vehicle traction motor control circuit as claimed in claim 2 in which said sample and hold circuit has a control terminal connected to the firing control means so as to cause sampling of the amplifier outlet in an interval between firing of the third thyristor and firing of the second thyristor.
 4. An electric vehicle traction motor control circuit as claimed in claim 3 in which a delay circuit is provided which is connected to the third thyristor firing means for causing said sample and hold circuit to sample the amplifier output a fixed delay after firing of the third thyristor.
 5. An electric vehicle traction motor control circuit as claimed in claim 1 in which said driver operable control means includes a current demand signal generator and said first and second firing control means includes a common comparator having a feedback circuit whereby said upper and lower limits are established, said comparator receiving the demand signal and a signal corresponding to the actual motor current and said commutating capacitor voltage sensing means acting to overringly limit the demand signal to a level determined by the commutating capacitor voltage.
 6. An electric vehicle traction motor control circuit as claimed in claim 5 in which said current demand signal generator is connected to the comparator through the intermediary of a circuit including an operational amplifier, a pair of series resistors connecting the output terminal of the operational amplifier to the base of an emitter follower transistor the emitter of which is connected to the comparator, a capacitor connecting the junction of the series resistor to a supply rail and a feedback resistor connecting the emitter of the emitter follower transistor to the inverting input terminal of the operational amplifier, the commutating capacitor voltage sensitive means including a diode connected to the base of said emitter follower transistor whereby said means can reduce the base voltage of the emitter follower transistor, but not increase such base voltage. 